Re: Ideas on causes of excess noise in transimpedance pre-amp?
- From: bill.sloman@xxxxxxxx
- Date: 20 Mar 2006 04:30:23 -0800
Bret Cannon wrote:
I'm trying to understand the noise I measure in a couple of versions of a
trans-impedance pre-amp that I'm using with a quartz tuning fork, such as is
used in digital watches, but removed from its can and used to measure
acoustic signals from a laser beam focused between the tines of the tuning
fork. The output noise spectra do not match what I calculate using the
Johnson noise of the feedback resistor, the specified voltage and current
noise of the op-amps, the measured feedback capacitance and my estimate of
the capacitance at the input of the op-amp.
The initial version of the trans-impedance amp used a Burr-Brown OPA656 with
a 10 Mohm feedback resistor on a 2-sided board. I measure an output noise
of 410 nV/rt-Hz at frequencies below 1 kHz, which matches the Johnson noise
of the 10 Mohm feedback resistor, but the noise floor starts increasing
above 1 kHz.
The corner at 1kHz implies a capacitance to ground of 16pF - a bit more
than your estimated 9.2pF.Printed circuit tracks are microstrip
transmission lines, and the capacitance per unit length depends on the
width. 50R tracks are fairly wide and run at 3.8pF/inch (150pF/metre).
75R tracks look like regular trances and run at 1.7pF/inch.
Philips use 2.2pF/inch in one of their application notes on using high
speed logic.
The extra 6pF suggests that you have neglected to allow for some three
inches of track over ground somewhere in your system.
This increasing output noise floor is qualitatively similar to
that due to the voltage noise of the op-amp amplified by the noise gain.
However the numbers don't workout. The specified input capacitance of the
OPA656 is 4.5 pF common mode, 0.7 pF differential. There are the
capacitances of the 1/2" long traces from the op-amp inputs to an SMA
connector soldered to the board, the SMA connector, an SMA to BNC adapter,
and a bulkhead BNC connector to which the fork is soldered. I estimate
these connector capacitances total about 3 pF. Adding in the shunt
capacitance of the quartz tuning fork, about 1 pF, the I estimate the total
input capacitance to be 9.2 pF. I measure a 3 dB bandwidth of 106 kHz from
which I conclude the stray capacitance of surface mount feedback resistor is
0.15 pF. With these capacitances and the 10M feedback resistor, I calculate
that the output noise should not rise above the Johnson noise of the
feedback resistor by 5% until about 100 kHz.
That sounds wrong. If the gain corner is at 1kHz, the noise gain at
30kHz is 30, so the 7nV/root Hz input noise of the op amp will look
like 210nV/root Hz at the output at 30kHz, 700nV/Hz at 100kHz
However I measure the noise reaching 500 nV/rt-Hz at 30 kHz. (With a quartz
tuning fork connected that is still in its can, there is a noise peak at the
fork resonance of 32764 Hz, but this noise peak has a width of 0.4 Hz and
doesn't not contribute significantly to the noise spectrum except within
about 20 Hz of the resonance frequency.)
The root means square sum of 210nV/root Hz and 400nV/root Hz is
452nV/root Hz.
The noise spectrum of the amplified inut noise won't be white, because
the high frequency components are amplified more than the low, so rms
summation may be over-optimistic.
This pre-amp has another problem, the trans-impedance gain varies with the
equivalent series resistance of the tuning fork's equivalent series RLC
circuit. The fork series resistance changes with pressure and temperature,
so the system gain changes with gas pressure. The change in trans-impedance
gain with source resistance is well fit by a model using a finite open-loop
gain of 62 dB for the OPA656 which is in fair agreement with the "typical"
open loop gain of 65 dB up to about 100 kHz.
So to reduce the variation in trans-impedance gain with source resistance, I
decided to replace the OPA656 with the OPA657. The OPA657 has a higher
typical open loop gain (75 dB) and a larger gain bandwidth product, but the
same specified input capacitance and resistance as the OPA656, the same
current noise (1.3 fA/rt-Hz), and slightly lower voltage noise (5 nV/rt-Hz
for the OPA657 and 7 nV/rt-Hz for the OPA 656). To my surprise and
disappointment, with the OPA 657, the noise increased at 30 kHz from 500
nV/rt-Hz to 600 nV/rt-Hz and peaks at 50 kHz. With the OPA656, the noise
kept increasing to 100 kHz, the upper limit of my spectrum analyzer.
To further confuse me, another version of this pre-amp was made with a
4-layer board that has provision for doing active bandpass filtering. We
found that the trans-impedance input stage with an OPA657 has twice the
noise at 30 kHz as the previous board with an OPA657. This higher noise
level is present when the components for filtering are removed. Even when a
quartz tuning fork is soldered to the 4-layer circuit board, the noise level
near 30 kHz is still about 3 times the Johnson noise level of the feedback
resistor.
I have measured the noise spectrum at the power pins for the OPA657 and it
is flat at 200 nV/rt-Hz from about 128 Hz until it starts rolling off at
about 40 kHz. I have also looked at the output of these pre-amps with a 400
MHz bandwidth scope and can not find evidence of oscillation. I found a
problem with current leakage on one of these boards that was fixed by
scrubbing with isopropyl alcohol and cotton swabs, so we've periodically
cleaned these boards without any reduction in noise since that first time.
The one other obvious difference in the two circuit boards is that on the
4-layer board, the ground plane extends under the op-amp and perhaps under
the traces from the op-amp inverting input to the fork connections, while
the original 2-layer board does not have a ground plane under the op-amp and
the inverting input.
My basic questions are:
1) What could be the cause of the excess noise that became significantly
worse when switching to a faster and higher gain op-amp that has less
voltage noise and that depends on the layout of the circuit board?
The stray capacitances on the printed circuit board seem to be higher
than you have estimated. Have you tried measuring the stray capacitance
on an unpopulated printed circuit board?
2) What things should be done in the layout of the next circuit board to
minimize the noise near 32 kHz for a trans-impedance pre-amp with gains of 1
to 10 Mohms?
Minimise the stray capacitance - shorter tracks, narrower tracks,
remove ground plane under the crucial tracks, as recommended in the
Burr-Brown application notes. Using a two-layer layout on the outer
layers of a four layer board reduces the distance to the ground plane
by a factor of three, and roughly triples the capacitance of any
microstrip tracks, even before you add in any extra ground plane.
Make sure that you have both the recommended 100nf decoupling capacitor
to ground close to the op amp's power pins, and a couple of uF of
tantalum capacitorless than an inch (25mm) away, as recommended by the
application notes. I tend to throw in a ferrite chip/bead between the
capacitors and the power rail - they are fairly cheap, don't introduce
any DC drop, and - being lossy non-wound devices - look resistive
rather than capacitative up to very high frequencies. The of the order
of 1uH of inductance can resonate with the decoupling capacitor, but as
long as you have a few uF of tantalum capacitor present, the resonance
is around 100kHz and the impedance at resonance low enough - at around
0R6 - that the ESR of the tantalum bead will kill any peaking.
--
Bill Sloman, Nijmegen
.
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